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Tri path Technol og y, I nc. - Techni cal I nformation
TCA2002 STEREO CLASS-TTM AUDIO CONTROLLER USING DIGITAL POWER PROCESSING (DPPTM) TECHNOLOGY
PRELIMINARY INFORMATION Revision 0.95 - April 2006
GENERAL DESCRIPTION
The TCA2002 is a two channel audio controller that uses Tripath's proprietary Digital Power Processing (DPPTM) technology. When combined with switching power output stages, the TCA2002 allows the implementation of a complete Class-T audio amplifier. Class-T amplifiers offer both the audio fidelity of Class-AB and the power efficiency of Class-D amplifiers. The TCA2002 is pin compatible with the Tripath's previous stereo controllers, the TC2000 and TC2001. The basis for Tripath controllers is a proprietary, fully scalable, feedback structure. The TCA2002 is capable of single ended or bridged operation, with single or split power supplies. When mated with the appropriate power stage, amplifiers with output powers of 25W to 1000W, or more, are possible. Both Tripath power stages as well as third party power stages, whether integrated, or discrete, can be used to make a complete, high-performance switching amplifier. Many features on the TCA2002 have been improved including increased supply range, elimination of turn-on pop, and automatic recovery from an overcurrent fault.
APPLICATIONS FEATURES
High End Amplifiers Professional Audio Amplifiers 5.1-Channel DVD Mini/Micro Component Systems
BENEFITS
High fidelity, high efficiency Class-T controller Feedback structure allows usage of unregulated power supply Analog inputs Improved Click and Pop performance Wider power supply range
Class-T Architecture Audiophile Quality performance when mated with appropriate power circuitry Supply range configurable via external resistors Wide Dynamic Range >100dB Compatible with unregulated and regulated power supplies Break Before Make (BBM) circuitry Over voltage and Under voltage circuitry with wider supply range Fully deglitched comparators for overcurrent detection Automatic recovery from fault conditions Pin compatible with TC2000 and TC2001
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A B S O L U T E M A X I M U M R A T I N G S (Note 1)
SYMBOL V5 Vlogic TA TSTORE TJMAX ESDHB ESDMM 5V Power Supply Input Logic Level
PARAMETER
Value 6 -0.3 to V5 +0.3 -40 to 85 -55 to 150 150 2000 200
UNITS V V C C C V V
Operating Free-air Temperature Range Storage Temperature Range Maximum Junction Temperature ESD Susceptibility - Human Body Model (Note 2) ESD Susceptibility - Machine Model (Note 3)
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. See the table below for Operating Conditions. Note 2: Human body model, 100pF discharged through a 1.5k resistor. Note 3: Machine model, 220pF - 240pF discharged through all pins.
O P E R A T I N G C O N D I T I O N S (Note 4)
SYMBOL V5 TA Supply Voltage Operating Temperature Range PARAMETER MIN. 4.5 -40 TYP. 5 25 MAX. 5.5 85 UNITS V C
Note 4: Recommended Operating Conditions indicate conditions for which the device is functional. See Electrical Characteristics for guaranteed specific performance limits. THERMAL CHARACTERISTICS
SYMBOL JA PARAMETER Junction-to-ambient Thermal Resistance (still air) Value 80 UNITS C/W
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E L E C T R I C A L C H A R A C T E R I S T I C S (Note 5)
TA = 25 C. See Application/Test Circuit.
SYMBOL Iq IMUTE VIH VIL VOH VOL VOH VOL VTOC IVPPSENSE PARAMETER Quiescent Current (Mute = 0V) Mute Supply Current (Mute = 5V) High-level input voltage (MUTE) Low-level input voltage (MUTE) High-level output voltage (HMUTE) IOH = XmA Low-level output voltage (HMUTE) Low-level output voltage (Yx, YxB) Over Current Comparator Voltage Threshold (OCD1 and OCD2) VPPSENSE Threshold Currents IOL = XmA TBD TBD 0.85 Over-voltage turn on (muted) Over-voltage turn off (mute off) Under-voltage turn off (mute off) Under-voltage turn on (muted) Over-voltage turn on (muted) Over-voltage turn off (mute off) Under-voltage turn off (mute off) Under-voltage turn on (muted) Over-voltage turn on (muted) Over-voltage turn off (mute off) Under-voltage turn off (mute off) Under-voltage turn on (muted) Over-voltage turn on (muted) Over-voltage turn off (mute off) Under-voltage turn off (mute off) Under-voltage turn on (muted) 121.6 61.2 45.6 58.0 46.6 23.4 17.5 153.4 118.7 58.3 41.0 60.1 46.5 22.9 16.1 0.97 1.09 151.4 IOL = XmA High-level output voltage (Yx, YxB) IOH = XmA TBD TBD V5 = 5V V5 = 5V 3.5 1.0 CONDITIONS MIN. TYP. 45 TBD MAX. 60 TBD UNITS mA mA V V V V V V V A A A A V V V V A A A A V V V V
VVPPSENSE Threshold Voltages with RVPP1 = RVPP2 = 383K (Note 6) IVNNSENSE VNNSENSE Threshold Currents
VVNNSENSE Threshold Voltages with RVNN1 = 392K RVNN2 = 1.2M (Note 6)
Note 5: Minimum and maximum limits are guaranteed but may not be 100% tested. Note 6: These supply voltages are calculated using the IVPPSENSE and IVNNSENSE values shown in the Electrical Characteristics table. They are shown for example only and are based on a +/-45V typical supply voltage. The voltage values shown are calculated using RVPP and RVNN values without any tolerance variation. Tripath recommends 1% tolerance (or better) sense resistors for all applications. Please refer to the Application Information section for a more detailed description of how to calculate the over and under voltage trip voltages for a given resistor value.
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TCA2002 PIN DESCRIPTION Pin 1 2 6 3 4 7 5 8 9, 12 10, 11 13 14,16 15 17 18 19 20 21 22 27 23, 28 24 25, 26 Function BIASCAP FBKP2 FBKP1 DCMP FBKN2 FBKN1 VPWR HMUTE Y1, Y2 Y1B, Y2B NC OCD2, OCD1 REF VNNSENSE OVRLDB VPPSENSE AGND V5 OAOUT1 OAOUT2 INV1, INV2 MUTE BBM1, BBM0 Description Bandgap reference times two (typically 2.5VDC). Used to set the common mode voltage for the input op amps. This pin is not capable of driving external circuitry. Positive switching feedback (Channels 2 & 1) Internal mode selection. Grounding this pin enables 16-pulses during startup. Connecting this pin to V5 disables 16-pulses during startup, which minimizes turn-on pop. DCMP must be grounded when connected to fet drivers or power stages using bootstrapped high-side supplies. Negative switching feedback (Channels 2 & 1) Test pin. Must be left floating. Logic output. A logic high indicates both amplifiers are muted, due to the mute pin state, or a "fault". Non-inverted switching modulator outputs. (Channels 1 & 2) Inverted switching modulator outputs. (Channels 1 & 2) No connect Over Current Detect inputs. If either pin exceeds VTOC, both amplifiers are muted. Ground if not used. The TCA2002 will try to automatically recover from an over current fault with an approximately 950mS repetition rate. Internal reference voltage; approximately 1.0 VDC. Connect and 8.25k, 1% to AGND. Negative supply voltage sense input. This pin is used for both over and under voltage sensing for the VNN supply. A logic low output indicates the input signal has overloaded the amplifier. Positive supply voltage sense input. This pin is used for both over and under voltage sensing for the VPP supply. Ground 5 Volt power supply input. Input stage output pins. (Channels 1 and 2) Single-ended inputs. Inputs are a "virtual" ground of an inverting opamp with approximately 2.5VDC bias. The bias at INV1 and INV2 is active, even if the MUTE pin is high. When set to logic high, both amplifiers are muted and in idle mode. When low (grounded), both amplifiers are fully operational. If left floating, the device stays in the mute mode. Ground if not used. Break-before-make timing control to prevent shoot-through in the output MOSFETs. Please refer to the Application Information section additional information.
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TCA2002 PINOUT
BIASCAP FBKP2 DCMP FBKN2 VPWR FBKP1 FBKN1 HMUTE Y1 Y1B Y2B Y2 NC OCD2
1 2 3 4 5 6 7 8 9 10 11 12 13 14
28 27 26 25 24 23 22 21 20 19 18 17 16 15
INV2 OAOUT2 BBM0 BBM1 MUTE INV1 OAOUT1 V5 AGND VPPSENSE OVRLDB VNNSENSE OCD1 REF
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APPLICATION / TEST DIAGRAM
TCA2002 Typical Application Circuit
RREF 8.25k, 1%
15 16
REF OCD1
OCD2 NC Y2 Y2B Y1B Y1
14 13 12 11 10 9 8 7 Y2 Y2B Y1B Y1 HMUTE RFBA 1K *RFBC 10K V5 COCR 220pF ROCR 20k
OCD2
OCD1 COCR 220pF ROCR 20k
VNN V5 V5 ROFB 510k V5 COF 0.1uF ROFB 510k CI 2.2uF IN1 V5 ROFB ROFA 10k 510k RF 20k
*RVNN1
17
HMUTE VNNSENSE FBKOUT1
OUT1 V5 *RFBB 1.1K RFBA 1K
*RVNN2 VPP V5 *RVPP2 CS 0.1uF *RVPP1
18
OVRLDB
CFB
19
VPPSENSE TCA2002
ROFA 10k
21
V5
FBKGND1
6
*RFBC 10K PGND1
20 AGND 22 23 OAOUT1 INV1 VPWR 5 4
*RFBB 1.1K
V5 RFBA 1K
+
RI 20k 1 2 24 MUTE
*RFBC 10K OUT2
FBKOUT2
25 COF 0.1uF CI 2.2uF
BBM1 V5
*RFBB 1.1K
CFB
ROFB 510k
RF 20k
26 27 28
BBM0 OAOUT2 INV2 DCMP FBKGND2 BIASCAP
3 2 1 CA 0.1uF
RFBA 1K
IN2
*RFBC 10K PGND2
+
RI 20k
*RFBB 1.1K
* Component values must be adjusted based on amplifier supply voltage
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E X T E R N A L C O M P O N E N T S D E S C R I P T I O N (Refer to the Application/Test Circuits)
Component RI RF CA CI RFBA RFBB RFBC
Description Inverting input resistance to provide AC gain in conjunction with RF. This input is biased at the BIASCAP voltage (approximately 2.5VDC). Feedback resistor to set AC gain in conjunction with RI. Please refer to the Amplifier Gain paragraph, in the Application Information section. BIASCAP decoupling capacitor. Should be located close to pin 1 and grounded at pin 20. AC input coupling capacitor, which, in conjunction with RI, forms a high pass filter at fC = 1 (2RICI ) . Feedback resistor divider connected to 5V. This resistor is normally set to 1k. Feedback divider resistor connected to AGND. The value of this resistor depends on the supply voltage setting and helps set gain in conjunction with RI, RF, RFBA ,RFBB, and RFBC. Feedback resistor connected from either the OUT1A/OUT2A to FBKOUT1/FBKOUT2 or OUT1B/OUT2B to FBKGND1/FBKGND2. The value of this resistor depends on the supply voltage setting and helps set gain in conjunction with RI, RF, RFBBA, RFBB, and RFBC. It should be noted that the resistor from OUT1/OUT2 to FBKOUT1/FBKOUT2 must have a power rating of greater than PDISS = VPP2 (2RFBC) . Feedback delay capacitor that both lowers the idle switching frequency and filters very high frequency noise from the feedback signal, which improves amplifier performance. The value of CFB should be offset between channel 1 and channel 2 so that the idle switching difference is greater than 40kHz. Potentiometer used to manually trim the DC offset on the speaker output. Resistors that limit the manual DC offset trim range and allows for more precise adjustment. Supply decoupling for the offset trim circuit. Bias resistor. Locate close to pin 15 and ground at pin 20. Supply decoupling for the power supply pins. For optimum performance, these components should be located close to the TCA2002 and returned to their respective ground as shown in the Application/Test Circuit. Main over-voltage and under-voltage sense resistor for the negative supply (VNN). Secondary over-voltage and under-voltage sense resistor for the negative supply (VNN). This resistor accounts for the internal VNNSENSE bias of 1.25V. Nominal resistor value should be three times that of RVNN1. Main over-voltage and under-voltage sense resistor for the positive supply (VPP). Secondary over-voltage and under-voltage sense resistor for the positive supply (VNN). This resistor accounts for the internal VPPSENSE bias of 2.5V. Nominal resistor value should be equal to that of RVPP1.
CFB
ROFA ROFB COF RREF CS RVNN1 RVNN2 RVPP1 RVPP2
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APPLICATION INFORMATION TCA2002 Basic Operation
The TCA2002 is a 5V CMOS stereo signal processor that amplifies the audio input signal and converts the audio signal to a 1-bt complementary, Class-TTM, switching pattern. This switching pattern is spread spectrum with a typical idle switching frequency of about 650kHz, externally adjustable by the output stage propagation delay and CFB. The switching patterns for the two channels are not synchronized and the idle switching frequencies should differ by at least 40kHz to avoid increasing the audio band noise floor. The idle frequency difference can be accomplished by offsetting the value of CFB for each channel. Complementary copies of the switching pattern are output through the Y1 and Y1B pins, and the Y2 and Y2B pins. These signals are used to drive the inputs of a MOSFET driver or integrated power stage. Additionally, "break-before-make" or dead-time can be added between the Yx and YxB transitions to minimize shoot-through currents in the output stage. This is adjusted by the BBM1 and BBM0 pins. The TCA2002 can be used over a wide range of supply voltages. The supply range is determined by the values of the feedback resistors in conjunction with the over-voltage / under-voltage sense resistors. The TCA2002 has over-voltage and under-voltage detection as well as overcurrent detection. These external inputs cause both channels of the amplifier to mute during the fault condition.
Class T Amplifier Architecture
TCA2002
FET Driver
MOSFETS
OUTPUT FILTER
Figure 1: Amplifier Architecture (One Channel) A basic Class T Amplifier Architecture is shown in Figure 1. There are three main blocks in the signal chain, though in some applications the FET Driver and MOSFETs may be integrated into a single package. The TCA2002 accepts an analog input signal. It processes this analog signal and converts it into a complementary bitstream. This bitstream serves as the input to the FET Driver. The FET Driver completes the appropriate level shifting and signal buffering to drive the gates of the Mosfet output stage. Differential feedback is taken at the output and is used by the TCA2002 to ensure high signal fidelity. The output filter recovers the amplified signal and connects to the loudspeaker. Figure 1 is a very simplified architecture diagram. Please refer to the more detailed block diagrams for the various amplifier configurations in the subsequent pages.
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Amplifier Gain
The gain of an amplifier based on the TCA2002 is the product of the input stage gain and the modulator gain. Please refer to the sections, Input Stage Design, and Modulator Feedback Design, for a complete explanation of how to determine the external component values for the various amplifier configurations.
A VTK2150 = A VINPUTSTAG
E
* A V MODULATOR
For example, using a TCA2002 with the following external components, in a Single Ended, Split Supply Application RI = 20k RF = 30.1k RFBA = 1k RFBB = 1.1k RFBC = 10.0k
AVTCA2002 -
20k 10.0k * (1.0k + 1.1k ) V + 1 = - 13.35 30.1k 1.0k * 1.1k V
Input Stage Design
The TCA2002 input stage is configured as an inverting amplifier, allowing the system designer flexibility in setting the input stage gain and frequency response. Figure 2 shows a typical application where the input stage is a constant gain inverting amplifier. The input stage gain should be set so that the maximum input signal level will drive the input stage output to 4Vpp.
TCA2002
OAOUT1 RF RI INV1 V5
CI INPUT1
+
BIASCAP
AGND V5
+ INV2 -
CI INPUT2
RF RI OAOUT2 AGND
Figure 2: Input Stage The gain of the input stage, above the low frequency high pass filter point, is that of a simple inverting amplifier: It should be noted that the input amplifiers are biased at approximately 2.5VDC. Thus, the polarity of CI must be followed as shown in Figure 2 for a standard ground referenced input signal
AV
INPUT STAGE
=-
RF RI
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Input Capacitor Selection
CI can be calculated once a value for RI has been determined. CI and RI determine the input low frequency pole. Typically this pole is set below 10Hz. CI is calculated according to:
CI =
where:
1 2 f P R I
RI = Input resistor value in ohms. fP = Input low frequency pole (typically 10Hz or below)
Modulator Feedback Design
The modulator feedback converts the high-voltage output-switching signal to voltage levels that the TCA2002 can process. The optimum gain of the modulator is determined from the maximum allowable feedback level for the modulator and maximum supply voltage for the power stage. Depending on the maximum supply voltage, the feedback ratio will need to be adjusted for proper operation and to maximize performance. The values of RFBA, RFBB and RFBC (see explanation below) define the gain of the modulator. In addition, the amplifier architecture affects the value of the feedback resistor values. Once these values are chosen, based on the maximum supply voltage, the gain of the modulator will be fixed even as the supply voltage changes. Thus, an amplifier based on TCA2002 has inherent PSRR unlike many other commercially available switching amplifiers. For the best signal-to-noise ratio and lowest distortion, the maximum differential modulator feedback voltage should be approximately 4Vpp. This will keep the gain of the modulator as low as possible and still allow headroom so that the feedback signal does not clip the modulator feedback stage.
Modulator Feedback Design
For SPLIT-SUPPLY, SINGLE ENDED operation:
VPP
TCA2002
FET Driver
MOSFETS
Y1 Y1B
HIN LIN
HS OUT LS OUT
HS GATE LS GATE
HB OUT
FBKP1
FBKN1
V5 R FBA R FBB R FBC R FBC
VNN
V5 R FBA R FBB
Figure 3: Split Supply, Single Ended Circuit (One Channel) Figure 3 shows the basic split supply, single ended amplifier using a TCA2002. The MOSFET output stage operates from two power supplies, VPP and VNN, that are equal magnitude, but opposite polarity (for instance +/-60VDC). One terminal of the load is driven by the half-bridge MOSFET output stage. The other load terminal is ground referenced. Differential feedback is taken from the
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half-bridge output and load ground. The resistor divider networks at the TCA2002 scale the feedback signals. The resistor values are determined based on the supply voltage, as explained below. The modulator feedback resistors are:
RFBA = User specified, typically 1K R FBA * VPP R FBB = (VPP - 4) R FBA * VPP R FBC = 4 R FBC * (R FBA + R FBB ) A V - MODULATOR +1 R FBA * R FBB
The above equations assume that VPP=|VNN|. For example, in a system with a SPLIT-SUPPLY, SINGLE ENDED amplifier of VPPMAX=60V and VNNMAX=-60V, RFBA = 1k, 1% RFBB = 1.071k, use 1.07k, 1% RFBC = 15.0k, use 15.0k, 1% The resultant modulator gain is:
AV - MODULATOR 15.0k * (1.0k + 1.07k ) + 1 = 30.02V/V 1.0k * 1.07k
For SPLIT-SUPPLY, BRIDGED OUTPUT operation:
VPP
TCA2002
FET Driver
HIN1 LIN1 HS OUT1 LS OUT1
MOSFETS
HS GATE1 HB OUTP LS GATE1
Y1 Y1B
HIN2 LIN2 FBKP1 FBKN1
HS OUT2 LS OUT2
HS GATE2
LS GATE2 HB OUTN
V5 R FBA R FBB R FBC R FBC
VNN
V5 R FBA R FBB
Figure 4: Split Supply, Bridged Output (One Channel) Figure 4 shows the basic split supply, bridged output amplifier using a TCA2002. The MOSFET output stage operates from two power supplies, VPP and VNN, that are equal magnitude, but opposite polarity (for instance +/-60VDC). Both terminals of the speaker are actively driven. Differential feedback is taken from each half-bridge output. The resistor divider networks at the TCA2002 scale the feedback signals. The resistor values are determined based on the supply voltage, as explained below.
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It should be noted that only one channel of the modulator is required to create a bridged amplifier. This is done by reversing the Y1 and Y1B connections to the second FET driver. This creates an out of phase signal at HB OUTN with respect to HB OUTP. A bridged amplifier has the advantage of potentially 4 times the output power for a given supply rail. But the disadvantage is that the required number of FET drivers and MOSFETs per channel increases by a factor of two. The modulator feedback resistors are:
RFBA = User specified, typically 1K R FBA * VPP R FBB = (VPP - 2) R FBA * VPP R FBC = 2 R FBC * (R FBA + R FBB ) A V - MODULATOR +1 R FBA * R FBB
The above equations assume that VPP=|VNN|. For example, in a system with a SPLIT-SUPPLY, BRIDGED OUTPUT amplifier of VPPMAX=60V and VNNMAX=-60V, RFBA = 1k, 1% RFBB = 1.071k, use 1.07k, 1% RFBC = 15.0k, use 15.0k, 1% The resultant modulator gain is:
AV - MODULATOR 15.0k * (1.0k + 1.07k ) + 1 = 30.02V/V 1.0k * 1.07k
For SINGLE-SUPPLY, BRIDGED OUTPUT operation:
VPP
TCA2002
FET Driver
HIN1 LIN1 HS OUT1 LS OUT1
MOSFETS
HS GATE1 HB OUTP LS GATE1
Y1 Y1B
HIN2 LIN2 FBKP1 FBKN1
HS OUT2 LS OUT2
HS GATE2
LS GATE2 HB OUTN
V5 R FBA R FBB R FBC R FBC
V5 R FBA R FBB
Figure 5: Single Supply, Bridged Output (One Channel)
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Figure 5 shows the basic single supply, bridged output amplifier using a TCA2002. The MOSFET output stage operates from a single power supply, VPP (for instance 60VDC). Both terminals of the speaker are actively driven. Differential feedback is taken from each half-bridge output. The resistor divider networks at the TCA2002 scale the feedback signals. The resistor values are determined based on the supply voltage, as explained below. It should be noted that only one channel of the modulator is required to create a bridged amplifier. This is done by reversing the Y1 and Y1B connections to the second FET driver. This creates an out of phase signal at HB OUTN with respect to HB OUTP. A bridged amplifier has the advantage of potentially 4 times the output power for a given supply rail. But the disadvantage is that the required number of FET drivers and MOSFETs per channel increases by a factor of two. The modulator feedback resistors are:
R FBB = User specified, typically 1K
R FBC = 350 * VPP - 1000 2333.33 * R FBC (1000 + R FBC ) R FBC * (R FBA + R FBB ) A V - MODULATOR +1 R FBA * R FBB R FBB =
For example, in a system with a SINGLE-SUPPLY, BRIDGED OUTPUT amplifier of VPPMAX = 60V, RFBA = 2.22k, use 2.21k, 1% RFBB = 1k, 1% RFBC = 20k, 1% The resultant modulator gain is:
AV
- MODULATOR
20.0k * (1.0k + 2.21k ) + 1 = 30.05V/V 1.0k * 2.21k
Mute Control
When a logic high signal is supplied to MUTE, both amplifier channels are muted (both high- and lowside transistors are turned off). When a logic level low is supplied to MUTE, both amplifiers are fully operational. Please note, that unlike previous Tripath controllers such as TC2000 and TC2001, the state of MUTE does not effect the input stage biasing. Thus, even if MUTE is high, the bias at INV1 will be active. This will keep the input capacitor, CI, charged via RI, thus, minimizing turn-on pops. Please note that the TCA2002 requires about 250mS to become active, after the de-assertion of MUTE (MUTE logic high MUTE logic low). Also, the TCA2002 requires about 350uS to go into mute, after the assertion of MUTE (MUTE logic low logic high).
Break-Before-Make (BBM) Timing Control
The TCA2002 can also insert a delay between the Yx and YxB output signals. This dead time will minimize shoot thru currents, in the case that the output driver or power stage does not have built in BBM circuitry. BBM0 and BBM1 are logic inputs (connected to logic high or pulled down to logic low) that control the break-before-make timing of the output transistors according to the following table.
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BBM1 0 0 1 1
BBM0 0 1 0 1
Delay 120 ns 80 ns 40 ns 0 ns
Table 1: BBM Delay The tradeoff involved in making this setting is that as the delay is reduced, distortion levels improve but shoot-through and power dissipation increase. The actual amount of BBM required is dependent upon components such as MOSFET type and gate resistor value as well as circuit board layout. The BBM value selected should be verified in the actual application circuit board. It should also be verified under maximum temperature and power conditions since shoot-through in the output MOSFETs can increase under these conditions, possibly requiring a higher BBM setting than at room temperature. The voltage at the BBM pins is sampled at the falling edge of mute. Thus, the BBM cannot be dynamically adjusted based on the output level of the amplifier.
Output Voltage Offset
The TCA2002 does not have internal compensation for DC offset. The circuit shown in the Typical Application circuit is a simple passive circuit that covers the range of expected offset voltage. Tripath has had success with both active and passive circuits for this purpose; please consult with the Tripath Applications team for further information.
HMUTE
The HMUTE pin is a 5V logic output that indicates various fault conditions within the device. HMUTE will be high if the MUTE input is connected to a logic high level.
OVRLDB
Also,
The OVRLDB pin is a 5V logic output that is asserted just at the onset of clipping. When low, it indicates that the level of the input signal has overloaded the amplifier resulting in increased distortion at the output. The OVRLDB signal can be used to control a distortion indicator light or LED through a simple buffer circuit, as the OVRLDB cannot drive an LED directly. There is a 20K resistor on chip in series with the OVRLDB output. If only 1 channel of the TCA2002 is used, the OVERLDB pin will be permanently low, since one modulator will not be switching. This can be alleviated if the unused (not connected to a power stage) modulator is run in self-feedback mode, via a few external resistors. Please contact your local sales office for more information.
Over-current Protection
The TCA2002 has two over-current protection comparator inputs, located at OCD1 and OCD2. These inputs can be connected to an external power stage, over current circuit output, such as that of TP2150B or similar. When the voltage across ROCR becomes greater than VTOC (approximately 0.97V) the TCA2002 will mute. Please note that the voltage at a given OCDx pin will need to exceed VTOC for 3uS for the fault to register. The low pass filter formed by ROCR and COCR further enhances this time-based deglitching. Upon an over-current fault, the HMUTE signal will be pulled high and both Yx and YxB will be low. Please note that both channels are muted if an over-current condition occurs. The reaction to an over-current fault is different than previous Tripath controllers. Unlike previous controllers that latched an over-current fault, the TCA2002 will automatically try to recover. Once an
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over current fault has occurred, the HMUTE will be pulled high for approximately 700mS. At this point, HMUTE will go low, and the TCA2002 will begin its normal un-muting sequence for 250mS. Thus, the typical retry time from an over-current fault is 950mS If over-current detection is not needed, for example in the cases where the power stage has built in over-current protection, simply connect OCD0 and OCD1 to AGND. This will disable the over-current detection comparators in the TCA2002.
Over- and Under-Voltage Protection
The TCA2002 senses the power rails through external resistor networks connected to VNNSENSE and VPPSENSE. The over- and under-voltage limits are determined by the values of the resistors in the networks (see APPLICATION / TEST DIAGRAM). If the supply voltage falls outside the upper and lower limits determined by the resistor networks, the TCA2002 will mute. The HMUTE pin will be pulled high and the Yx and YxB pins will be low. The removal of the over-voltage, or under-voltage condition, returns the amplifier to normal operation. Please note that trip points specified in the Electrical Characteristics table are at 25C and may change over temperature. For applications where the TCA2002 is mated with a power stage that has built-in over and undervoltage protection, this protection feature can be disabled by connecting a 30kohm resistor from VPPSENSE (pin 19) to V5, and a 15k resistor from VNNSENSE (pin 17) to AGND. Please note that the MUTE pin must be held high until all power supplies are stable to eliminate any possible power supply sequencing problems. The TCA2002 has built-in over and under voltage protection for both the VPP and VNN supply rails. The nominal operating voltage will typically be chosen as the supply "center point." This allows the supply voltage to fluctuate, both above and below, the nominal supply voltage. VPPSENSE (pin 19) performs the over and undervoltage sensing for the positive supply, VPP. VNNSENSE (pin 17) performs the same function for the negative rail, VNN. When the current through RVPPSENSE (or RVNNSENSE) goes below or above the values shown in the ELECTRICAL CHARACTERISTICS section (caused by changing the power supply voltage), the amplifier will be muted. VPPSENSE is internally biased at 2.5V and VNNSENSE is biased at 1.25V. Once the supply comes back into the supply voltage operating range (as defined by the supply sense resistors), the amplifier will automatically be un-muted and will begin to amplify. There is a hysteresis range on both the VPPSENSE and VNNSENSE pins. If the amplifier is powered up in the hysteresis band the amplifier will be muted. Thus, the usable supply range is the difference between the overvoltage turn-off and under-voltage turn-off for both the VPP and VNN supplies. It should be noted that there is a timer of approximately 200mS with respect to the over and under voltage sensing circuit. Thus, the supply voltage must be outside of the user defined supply range for greater than 200mS for the amplifier to be muted. Figure 6 shows the proper connection for the Over / Under voltage sense circuit for both the VPPSENSE and VNNSENSE pins.
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TCA2002 - KLi/0.95/04.06
Tri path Technol og y, I nc. - Techni cal I nformation
V5
VNN
TCA2002
RVNN1
RVNN2
17
V5 VPP
VNNSENSE
RVPP2
RVPP1
19
VPPSENSE
Figure 6: Over / Under voltage sense circuit The equation for calculating RVPP1 is as follows:
R VPP1 =
VPP I VPPSENSE
Set R VPP2 = R VPP1 . The equation for calculating RVNNSENSE is as follows:
R VNN1 =
VNN I VNNSENSE
Set R VNN2 = 3 x R VNN1 . IVPPSENSE or IVNNSENSE can be any of the currents shown in the Electrical Characteristics table for VPPSENSE and VNNSENSE, respectively. The two resistors, RVPP2 and RVNN2 compensate for the internal bias points. Thus, RVPP1 and RVNN1 can be used for the direct calculation of the actual VPP and VNN trip voltages without considering the effect of RVPP2 and RVNN2. Using the resistor values from above, the actual minimum over voltage turn off points will be:
VPP MIN_OV_TUR VNN MIN_OV_TUR
N_OFF
= R VPP1 x I VPPSENSE (MIN_OV_TU RN_OFF) N_OFF = - ( R VNN1 x I VNNSENSE (MIN_OV_TU RN_OFF) )
The other three trip points can be calculated using the same formula but inserting the appropriate IVPPSENSE (or IVNNSENSE) current value. As stated earlier, the usable supply range is the difference between the minimum overvoltage turn off and maximum under voltage turn-off for both the VPP and VNN supplies.
VPP RANGE = VPP MIN_OV_TUR N_OFF - VPP MAX_UV_TUR N_OFF VNN RANGE = VNN MIN_OV_TUR N_OFF - VNN MAX_UV_TUR N_OFF
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TCA2002 - KLi/0.95/04.06
Tri path Technol og y, I nc. - Techni cal I nformation
Package Information - TCA2002
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TCA2002 - KLi/0.95/04.06
Tri path Technol og y, I nc. - Techni cal I nformation
PRELIMINARY INFORMATION - The above specification is for a device that is currently in development. All specifications and device descriptions are subject to change based on product analysis and characterization. Please contact Tripath Technology for additional information not included or covered in this advanced document. Tripath Technology Inc. reserves the right to make changes without further notice to any products herein to improve reliability, function or design. Tripath does not assume any liability arising out of the application or use of any product or circuit described herein; neither does it convey any license under its patent rights, nor the rights of others. Tripath and Digital Power Processing are trademarks of Tripath Technology Inc. Other trademarks referenced in this document are owned by their respective companies. TRIPATH'S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN CONSENT OF THE PRESIDENT OF TRIPATH TECHNOLOGY INC. As used herein: 1. Life support devices or systems are devices or systems which, (a) are intended for surgical implant into the body, or (b) support or sustain life, and whose failure to perform, when properly used in accordance with instructions for use provided in this labeling, can be reasonably expected to result in significant injury to the user. 2. A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or effectiveness. For more information on Tripath products, visit our web site at: www.tripath.com
Contact Information
TRIPATH TECHNOLOGY, INC 2560 Orchard Parkway, San Jose, CA 95131 408.750.3000 - P 408.750.3001 - F For more Sales Information, please visit us @ www.tripath.com/cont_s.htm For more Technical Information, please visit us @ www.tripath.com/data.htm
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TCA2002 - KLi/0.95/04.06

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